Monday, 9 November 2015

That 90 Degree Phase Shift


After much thought, I'll focus my further receiver efforts on analog domain phasing receiver design and construction. While great tools, superhet receivers lost their luster for me. Plus, I'll no longer have to deal with birdies, crystal filters, RF mixer products and the IF image.

Quadrature detectors with digital signal processing don't excite me either, however, that may change over time.

For over 40 years I've listened to detected RF through a speaker — enjoying mostly hi fidelity audio. In 2015, I find no reason to change my preferred listening practices. To that end, direct conversion, or zero-IF ( ZIF ) receivers deliver a sonic impact that I don't seem to get with my superhets. Further, digital processing may be applied along the ZIF receiver chain as I eventually modernize my analog dominant, primitive, hobby radio experiments.

Good DC receiver design poses many challenges; especially when you apply phase shifting techniques to suppress the unwanted side band by > 30 dB. Effectively reducing analog gain and/or phase imbalances between the translated I and Q baseband signals without relying on digital signal processing proves no small task. Unrelated issues such as DC offset due to local oscillator leakage also lurk.

How do we obtain a precise, wide band 90 degree phase shift at RF — and also at AF? Presented are some of my first experiments at making 90º analog phase shifts from AF to UHF.

As a 90 degree phase shift newbie, reading the material written by Rick, KK7B published in EMRFD Chapter 9 formed my inaugural task [ Reference #1 ]. The R2Pro /KK7B Designs Yahoo group also well supports the Chapter 9 and related material. Since modern cell phones and many other receivers apply ZIF I-Q techniques, we may also find numerous online resources to read.

[1]  RF Quadrature Hybrid

I started with the classic 7 MHz "Fisher" quadrature hybrid [ Reference # 2 ] also presented in EMRFD Chapter 9.

Above — We've seen this 3 dB quadrature hybrid schematic in many ARRL articles for 7 MHz.

Above — I quickly built the above 3 dB hybrid with no special attention towards matching the 2 capacitors, or even building a precise layout. I wanted to measure it as a 'typical' build that a beginner might ply. We're so use to Y/T scope graphics that this lovely, round, Lissajous curve from an X-Y plot jumps out at you. This 7 MHz twisted wire 3 dB hybrid coupler works like a charm.

Above — Some Y/T DSO analysis. I measured the insertion loss @ 3.02 dB at Fc. To classify the bandwidth I adopted what I saw on a few datasheets: +/- 10% of the center frequency. I determined my Fc @ 7.060 MHz and felt surprised with its performance.

Like what I've read in the literature, signal amplitude varies much more than phase between the 2 ports as you move up and down in frequency. It's 1 thing to read information  — and something entirely different to see it happening before you eyes during real bench experiments.

This simple quadrature will work for the whole 7 MHz Ham band in many phasing receivers.

Above — I wanted to see how far up I could take the twisted wire quadrature in frequency.  I
  1. Twisted 2 wires together.
  2. Smashed up a T30-12 toroid inside a plastic sandwich bag with a hammer.
  3. Made a slurry of glue + Fe material and dabbed it on the center of the twisted wire.
  4. Measured the L at 26.5 nH and calculated that XL = 50 Ω @ 602.4 MHz.
  5. Crudely built a tiny 3 dB hybrid with trimmer capacitors to get the needed XC of 100 Ω.

Above — DSO analysis showed that it worked!  Some brief experiments seemed to indicate that a reasonable bandwidth might be possible if careful UHF breadboard practices were applied. I chose Fe material from the T30-12 since it had the lowest permeability of any toroid in my collection. Perhaps, my next experiment should involve no ferrous material?

Commercial 90 º splitter / combiners are available for multiple frequencies. Click for 1 example.

Branch line UHF Quadrature Hybrid

I wanted to make 3 dB quadrature hybrid for ~435 - 438 MHz: the ~ 70 cm amateur satellite band.

Commonly, builders insert a 90º λ/4 transmission line delay to make a power divider. Although these lack port isolation, they're theoretically easy to do.

To explain, I’ve never enjoyed glad outcomes with cutting and fitting λ /4 coaxial transmission lines on indoor bench projects — the likely outcome = leave the lab. That’s out for me. I'm told that semi-rigid hard line is easier to work with, but the cost seems prohibitive.

Another commonly applied transmission line structure = the 2 branch quadrature hybrid with
transmission lines made from coax, strip line or microstrip.  Although appealing — for me, at least,  lower UHF = no person’s land because λ is too large to make practical size circuits.

In my opinion, microstripline transmission line techniques seem best suited at and above 2 GHz where the boards get reasonably small to fit into standard radio enclosures.

I thought about carving a classic 90 degree, λ/4 branch-line 3 dB coupler on a copper board, but, as mentioned, it’s too big for my liking. Another option includes making a reduced-size branch-line coupler with a capacitor tuning each arm's end. That’s what I did.

Above — My basic design. The reduced-size branch-line coupler offers lower bandwidth than  versions built with proper λ/4 branch-lines.

Above — My Ugly branch line coupler build. Hand carved outdoors with a motorized tool in the wet, cold, fall weather, it does not look too pretty. Function always trumps looks in my book. This was my second version and featured 2-sided FR-4 board with some removal of the top side ground plan around the branch line paths. I connected all top ground plane sections to the lower copper ground plane with copper via wires.

 Above — I initially tuned it at ~436 MHz with an X-Y plot on my DSO

Above — Analysis showed the best bandwidth and phase match occurred with the hybrid centered at 438.6 MHz — presumably due to measurement + cutting errors. Bandwidth was low, but I could tune the entire ~70 cm Amateur satellite band with reasonably tight amplitude and phase balance. At least I've got something to start experimenting with and — gained a little experience up at UHF.

[2] Audio Frequency 90 Degree Phase Shifting

Rick, KK7B and others wrote that the I - Q baseband phase + amplitude imbalances in our RF quadrature hybrid and down converting mixers +/- first AF amp output ports may serve as deal breakers for getting maximal opposite side band suppression in ZIF phasing receivers.

With further reading and thinking about this on my plate, I'll just focus on the op-amp, all-pass 90 º phase shifter block applied between the post mixer (+/- first AF amp) I + Q channels and the combiner.

To start with, I designed and built a simple, low bandwidth, add-on 90º phase shifter for my 1 KHz bench audio signal generators:

Above — The schematic for my 1 KHz 90º phase shifter. The 10 nF capacitors were 1% tolerance. I show the calculated resistor values, however, for other than the 10K gain/feedback resistors, my build resistors were standard value 5% types.

Likely over-designed, I quickly built this on a whimsy to get me going with all-pass filter design. In most all-pass filter work, you've got to design it — and then order and wait for your 1% resistors to arrive. No lag for me: I went from design to test in about 2.5 hours. Previous to the experiments on this page, I'd never even thought about all-pass networks — now I feel excited to learn about and work with them.

Above  — Assessment of my narrow band phase shifter. I got pretty close to the design phase error by sorting through my 5% resistors with an ohmmeter and choosing time constant Rs as close as possible to those specified in the schematic. I might order some nearest standard value 1% Rs and see if I can get under 0.1% phase accuracy @ 1 KHz.

Now, along with boosted confidence, I've got something to connect to my AF signal generator for simple assessment of the wide band all-pass networks I build.

 Above  — 2 more photographs of the 1 KHz phase shifter.

Wide band All-Pass Phase Shifters

Above  — The general form of wide band all-pass filters applied in analog phasing receivers.
The R-C time constant gets fairly critical if you want the maximum possible opposite side band suppression, so buying 1% parts ranks as important.

The most important task for me was to re-read EMRFD Chapter 9.  I can find no greater reference with both mathematical and experiential writing. From discussions in the R2Pro / KK7B Designs Yahoo group I learned why Rick chose an all-pass bandwidth of 270 to 3600 Hertz with 0.1% amplitude + phase error.

I won't repeat it, but getting the phase shifter bandwidth as wide ( & flat) as possible; plus comparing various bandwidth all-pass designs with an antenna + speaker attached = key learnings. Rick's filter presented as Figure 9.56 in EMRFD remains a proven, widely reproduced, go-to, all pass network for many.  I ordered all the 1% resistors yesterday.

It's also heavy to learn that all opposite side band suppression occurs in the op-amp combiner that immediately follows the all-pass network.

I wanted to try designing some wide band all-pass networks to learn more about them.

Above  — My single resource for AF filter design: Electronic Filter Design Handbook by Arthur B.Williams. [ Reference # 4 ]. That book and Handbook of Filter Synthesis by Анатоль Зверев (1967) serve as the archetype references for analog filter designers. Get them.

Although you can buy software to crunch the math and design wide band, all-pass filters, there is something so organic about looking at tables and grinding out maths with a scientific calculator.

From the Williams book, I designed some filters with various bandwidth from the α constants and method provided.
Above  — The design of a 250 - 3000 Hz all-pass filter that fits the 3 section per side, all-pass filter template shown earlier.  My raw R values were substituted with nearest standard 1% metal film resistor values from a Vishay Dale precision resistor decade table.

Ken Kuhn wrote an Excel spreadsheet that plots a graph based on the RC time constants at R1 to 6. He later improved the spread sheet so you can just enter resistor and capacitor values for 1 to 3 all-pass sections.  Thus you don't need to calculate the time constant as shown in my work.

Ken granted permission to share the spreadsheet. Click for his file.

 Above  — The graph of my filter shows too much slop below 2 KHz

Above  — The beauty of his spreadsheet = tweaking. I inputted my raw, calculated R1 - 6 values then tweaked R1 to get this lovely transfer function. I will try tweaking with 1% resistor values next.

2 weeks ago I didn't know anything about all-pass filters, now I'm able to at least mathematically design them and more importantly, understand a little about them.

The  Table 1  α1 - 6  constants will pretty much work at any reasonably wide bandwidth where you want to get a +/- 0.1 % phase error. You must use 6 total op-amp sections like the all-pass filter template shows.

I plan to experiment to learn more about ZIF receiver topics including the Weaver method of processing the I and Q channels. The material published by Matjaž, S53MV about his ZIF receivers plus his other designs serves as great inspiration to me.

My special thanks to Ken Kuhn for writing and sharing his spreadsheet; Rick, Allison and others on the R2Pro/KK7B Designs Yahoo group —  and to you for reading my blog.


[1]  Experimental Methods in RD Design (EMRFD) First published by the ARRL in 2003. Wes Hayward, W7ZOI, Rick Campbell, KK7B and Bob Larkin, W7PUA.

[2]  Twisted-Wire Quadrature Hybrid Directional Couplers for QST, January  1978. Reed Fisher, W2CQH.

[3]  Free TI Software to design op-amp filters. Click. I used this to design my 1 KHz all-pass filter.

[4]  Electronic Filter Design Handbook. Arthur B. Williams.  McGraw-Hill 1987.

[5]   From Thomas, LA3PNA : Sage wireline.  Basically, just 2 twisted pieces of wire in a piece of copper tubing. Some enamel wire in a thin KS brass tube should do the trick in a homebrew version. I believe it could be coiled up if the total length is to long in a given space, the important part should be that the tube is grounded at both ends.  Found this note about the length in ADS:  The quarter-wavelength frequency is calculated as:  F (MHz) = 1850 / L (inches). Click for wonderful instructional sheet with math + photos.

Thursday, 29 October 2015

QRP WorkBench Line-in Audio Amplifier — Part 2

Welcome to Part 2

I share some experiments, plus a few thoughts & observations from my QRP workbench.

[A]  Some general points in AF design and breadboarding

[1] Expect your amplifiers to oscillate and design to mitigate this.

[2] Read your datasheet(s).

[3] As possible, identify the frequency of any parasitic oscillations by measurement. If you lack a 'scope, you can’t hear HF oscillations, but may hear their outcomes — or view their effects on DC measures like quiescent current.

[4] Going for high gain in a single stage may bring you pain.

[5] Build a homebrew, low noise AF signal generator with an amplitude control pot — if only for 1 frequency such as 1 KHz. Your sure to benefit from the experience designing for and soldering on op-amps.

[6] Test your AF boards before inserting them in a receiver.

[B]  AF + HF Decoupling and Bypassing

For this discussion, I placed an LM386 in high gain mode ( 46 dB power gain ) and measured the output signal across an 8 Ω load resistor.

To show what can happen if power supply Pin 6 is unbypassed; or inadequately decoupled and bypassed @ AF,  I left off the usual Pin 6  R-C low-pass filter network. Vs = 12.3 VDC.

Above — 144.6 Hertz oscillations erupt with no input signal applied by my signal generator.

Insufficient power supply filtering allows low frequency noise to make it from the output into the input via the DC power line where that feedback path contributes 180 degrees of phase shift to the 180 degrees provided by the inverting amplifier.  If the gain at the oscillation frequency >=1  you'll hear (and see with a 'scope ) AF parasitic oscillations.

The quiescent current of an LM386 powered by 12 VDC ran  5.6 - 6.2 mA in my Lab. I measured the current with the parasitic oscillations seen above and the quiescent current rose to 13.2 mA. A higher than expected quiescent current may signal parasitic AF - HF oscillations in an AF stage.

Above — I put a 1 KHz signal into that LF oscillating LM386 input.

Above — An FFT of above.  Looks and sounds nasty!  F0 = 1.015 KHz.

Above — A 470 µF AF bypass electrolytic capacitor was soldered to Pin 6, plus a 10 Ω decoupling resistor inserted between the power rail and Pin 6. Further, I AF bypassed the B+ rail with 100 µF.

Above — The averaged noise voltage at the LM386 output after adding the DC line low-pass filtration shown above. Parasitic audio oscillations killed dead!

Above — 3 different DC power line filtering R-C networks set up for analysis in a 50 Ω system. In our AF amplifiers, we need to worry about parasitic HF oscillations in addition to AF parasitics.

Above — A tracking generator plus spectrum analyzer sweep of Figure A.  The analyzer doesn't measure below 9 KHz and unfortunately doesn't assess the 470 µF capacitor. However, from other experiments and calculations, the 470 µF plus the 0.1 µF give a good wideband bypass for AF to HF.

The reference sweep is the red horizontal line above [ 0 dB attenuation of the signal ]. When inserted between the TG + SA, the 0.1 µF cap exerts it's ultimate attenuation of the signal at 5.8 MHz ( -44.3 dB = peak attenuation).

Above — TG + SA sweep of Figure B. Now we've got a classic pi filter with shunt capacitor(s) on each end separated by a 10 Ω resistor.  Peak attenuation = 8.04 MHz. Notice how the pi filter deepens the ultimate attenuation of the 0.1 µF and gives better RF bypass than just a single capacitor with no decoupling resistor.
This rings true for both RF and the AF bypass capacitors.

Above — TG + SA sweep of Figure C.  R now = 100 Ω. The peak attenuation occurs at ~12.1 MHz and has flattened out even more to provide better RF bypass across the span. This is why we decouple and bypass the DC lines in our RF projects too.

Above — To get better DC filtering @ HF in AF amplifiers, we ought to put a at least 1 HF bypass capacitor on our positive supply rail to make the classic pi filter. In Figure A, I've placed AF + HF bypass on both sides of the 10 Ω decoupling resistor. We're radio builders and don't want parasitic AF or RF flowing down our B+ lines and wreaking havoc in our AF or RF stages.

For op-amp DC power supply pin AF bypass, 22 µF works well in my Lab, however, you've got to find what works best from your own experiments.  

At Figure B, the 10 Ω resistor is shown as optional. Since the power followers draw significant current for speaker-level volume during signal peaks, the voltage drop across the 10 Ω resistor might reduce clean signal power a little. I tend to leave off the 10 Ω resistor and only include it if I measure AF instability that I can't fix by increasing the value of the C within reason.

I've encountered AF parasitic oscillations with a 100 to 220 µF bypass capacitor in my power amplifier stage designs that was stopped by going up to 470 µF — now I just stick 470 µF in as my default AF bypass capacitor and work from there. On my bench, at least, higher gain PA stages tend to oscillate more. I strive to keep the power gain of my PA stages <= 26 dB.

Choosing gain, cap and resistor values are decisions we designers must face during every build. Consider thinking critically and choosing carefully — don't just copy what another builder did, because that builder might have copied someone else and so on. Some times unmeasured 'minimal part' designs that seem attractive dole out maximum grief and ruin your bench experience. Bench time should foster fun and discovery.

Above — A JavaScript applet screen shot from my blog  You've go to decide what R and C values to use for AF bypass and to decouple your DC power lines. I tend to ply lower Rs and higher C's to minimize DC voltage drops to preserve headroom. From the example above, if you applied a 100 Ω resistor, then only a 47 µF capacitor is required for the same 3 dB down frequency.

Yes, our parts collection often dictates what choices we make, but if you're buying parts — a  470 µF/25v capacitor often costs just pennies more than a 47 µF/25v capacitor — especially with low-cost parts offered by on-line stores and auctions.

[C] HF Oscillations 

Above — Intense HF oscillations from a 5532 op-amp circuit. I collected this and the other images over the past 2 years.

Above —  I didn't think this was possible — 10 MHz oscillations in an op-amp + power follower AF stage. The stage quiescent current measured ~ 65 mA just from these parasitic oscillations.

Above —HF oscillations in a LM4562 based op-amp tone circuit driven with a 1 KHz AF tone from my 1 of my homebrew audio signal generators.

Above — Techniques to remove HF oscillations. Decoupling and bypassing at HF previously discussed in Section [B]

Figure A — The input of any AF chain, or IC like the LM386 should contain RF bypass to ground. You might also choose that capacitor's value to shunt some of the higher frequency noise and signal to ground like a low-pass filter.

The output of the LM386, like most power amps, contains the familiar R-C ( Zobel ) network connected in parallel with the speaker voice coil. I recommend adding this to all PA output stages.

Since a speaker load presents a complex impedance, placing the Zobel network in parallel with the voice coil keeps the amplifier happy since it's a purely resistive load.  The resistive load boosts the PA's stability and in some high power, more sophisticated IC power amps, helps to eliminate negative voltages that could harm the PA.

Figure B — In high gain feedback amplifiers, it does not take much time delay, or phase lag to trigger high frequency oscillations near the upper end of the op-amp's bandwidth.  I've noticed 2 cases where this is more likely to occur: in op-amp driven PA stages, and also in some tone control circuits. I applied 20 pF for Cx in the tone circuitry applied in my line-in AF amp from Part 1 of this blog posting.

Figure C —  Similar to what's shown in Figure B; a small value feedback capacitor Cx lowers the closed loop bandwidth so there's insufficient gain at high frequencies for oscillation to occur. As aforementioned, in some stages you might create a frequency dependent situation where the total phase shift in the feedback loop exceeds 360 degrees and has gain larger than 1 which = unwanted oscillations.  Feedback capacitor Cy lowers the op-amps upper cutoff audio frequency, and as an added benefit, its noise bandwidth.

In typical complimentary emitter follower power amplifiers, this works well. An example = Figure 9.74 by Rick, KK7B shown in EMRFD. In my particular PA from QRP WorkBench Line-in Audio Amplifier — Part 1, too high a cap value will actually trigger HF oscillations.

[D] Environmental Noise

Noise sources may affect your measures. To conserve energy, I purchased an LED light bulb for my new workbench. I knew these things used switched drivers, but wanted to 'go green'. Bad mistake. I'll show you some cool DSO traces measured during some AF design work.

Above — I went to measure the noise voltage across an 8 Ω resistive output load of a PA with no input signal, plus a 470 Ω shunt resistor across the input. I could not figure out why my stage measured so crazy noisy!

Above — With some DSO averaging, this signal looked like my noise was modulated by a regular occurring oscillation of unknown cause.

Above — Finally, it occurred to me: it's probably the LED lamp above your work. I switched the LED bulb off and instantly the modulated noise stopped. Yikes!  I gave the bulb to a friend who lives across town.

Above — The LED bulb modulating an HF oscillating LM4562 op-amp circuit.

Above — That wretched LED bulb even modulated a low-level UHF circuit.  It's gone for good.

[E] LM386 Musings

Let's unpack the LM386 a little. It's a great design that finally went end-of-life this year. Over the years of web publishing AF stuff, this humble part provided me with many emails and many wanted to use it at 46 dB gain to eliminate the need to make a preamplifier — or as a space saver so they could stuff their entire project into a mint tin.

If you manage to build a high-gain mode LM386 AF amp without hum and parasitic oscillations, you may notice it gives some crunchy harmonic distortion when driven anywhere towards loud. My question = why?

I performed a ton of experiments and saved over 50 files. The tough part was writing something that made sense and stuck to the measures and facts. When we don't know facts, or information based on reliable measures and data, we often just get opinions : they're free on the Internet.

LM386 with Power Gain = 46 dB      "High gain mode"

Above — The best 2nd harmonic distortion possible even when driven to only 11.8 mW. In high gain mode, the harmonics do not clean up at low drive levels like other AF PA circuits I've tested.

Above — I drove this particular LM386 to 2 Vpp and 6.82 Vpp and we'll use those 2 voltages  to compare this to the chip with 26 dB gain (low gain mode). At 6.82 Vpp (or 727 mW) the sine wave starts to show obvious distortion, so I chose this as a benchmark since many builders don't have access to FFT.  The load = an 8 Ω resistor bank.

Above — Sine wave in time domain driven to 6.82 Vpp. I can just see the positive tip starting to distort. It's easier to do this live by bringing the waveform in and out of distortion repeatedly. From practice, I can usually see sine wave distortion when the 2nd harmonic lies >= -44 dBc — and especially when the 3rd harmonic moves towards this level.

Above —The FFT of the above sine wave. The third harmonic lies about -43 dBc.

LM386 with Power Gain = 26 dB

I set that LM386 to minimum gain mode by removing the capacitor between Pin 1 and 8. Lets see how it compares to the above measures @ 2 and 6.82 Vpp.

Above — The FFT when driven to 2 Vpp. Compared to the high gain mode, the 2nd and 3rd harmonics are ~ 7 dB and 5 dB lower respectively.

Above — The FFT driven to 6.83 Vpp. Compared to the high gain mode, the 2nd and 3rd harmonics are about 3 dB and 6 dB lower respectively.

From these 4 FFTs, I can't see why I seem to hear more distortion in the high gain mode version since the 2nd-3rd harmonic differences are <= 7 dB. Or, maybe that's enough of a difference to hear?

Perhaps it's just easier to overdrive a high gain LM386 in real life receiver testing? Consider to, I don't run AGC in my receivers, so my louder RF signals do sound louder. I'm pretty certain the distortion doesn't occur in my receiver front end, since I've heard it on peaks with attenuators switched in — and also in my high performance front-ends, plus non-radio projects.

Finally, here's an old experiment that makes no direct comparisons to any other LM386 configurations.

Above an  LM386 schematic and its FFT:  If I do run an LM386 AF PA in a receiver, I build a low noise (preferably op-amp) preamplifier and drive the LM386 in the so-called bass boost configuration as shown in the schematic just above the FFT.

An R-C network provides negative feedback from the output to the emitter of the input PNP transistor via Pin 1. I've noticed that in this particular configuration, as you lower the drive level, the harmonics clean up ( unlike what happens in the high gain mode and much better than in plain low gain mode ). In the above FFT, please view that all harmonics are 60 dBc down @ (and below) 235 mW output power.

I also bypass the point between the two 15K emitter resistors of the grounded PNP of differential pair @ Pin 7. This bypasses any DC line noise/ripple to ground.

Wait.  Now I'm getting it.  I don't just prefer the LM386 in low gain mode; I also prefer it with negative feedback. Good old negative feedback!

I'll let you draw your own conclusions about how and when to employ the venerable LM386. 

OK, enough ranting  — back to the bench!

Click for Part 1.

Sunday, 25 October 2015

QRP WorkBench Line-in Audio Amplifier — Part 1

Happy Fall  !

Each Fall, I view design media and experiment @ AF to boost my skills and know-how. A bigger bench means room for more test equipment & accessories. Lately, I'm enjoying a bench activity maxima.

In Part 1, I show a new line-in AF amplifier and in Part 2 some lessons learned about reducing noise and oscillations in home built AF circuitry.  Hopefully, you find these installments useful.

Part 1: Line-in Amp 

I make a lot of bench receivers and grew tired of repeatedly making new speaker-level AF amps. Now, on each receiver AF section, I'll just run a low-noise preamplifier, plus buffer stage(s) to raise the signal out of the noise and drive a low impedance output respectively. The in-situ buffer stage might also low-pass filter the signal before it leaves the receiver. A line-out RCA jack on the receiver chassis back will provide a connector for patching my receivers to this new AF amplifier box. Think modular approach.

I sought a low distortion, low noise, high fidelity project.

Above  —  Block out of the entire project along with the power gain of each stage.

Above  —  Outdoor photo.

Above  — The low impedance virtual ground (VG) source schematic and breadboard grounding system for this project.

Ground Paths

A carved square pathway lies around the 2 anchoring bolts for each of the 3 circuit boards so the bolts do not chassis ground these boards. A single wire connects each copper ground plane to a common point [ earth ground ] on the 2 DC voltage input/output jacks: I call this star grounding and used this technique when building 5 - 50 W tube guitar amplifiers years ago. I didn't isolate the input and output RCA jacks, although was prepared to if hum arose.

I won't go deeply into grounding since more informed people offer great content via an Internet search. The potential difference between 2 or more ground paths may induce a noise voltage source along those paths. When this happens, you'll hear a wretched buzz — 50 to 60 Hertz hum depending on which country you call home.

Star grounding removes alternate ground paths and thus reduces the potential for hum.
Still, others, isolate alternate ground paths with techniques including common mode chokes, isolation transformers and balanced circuitry. In typical homebrew receiver AF projects, ground loops don't pose the problem magnitude seen in larger AF projects such as a homebrew home audio system, or a tube guitar amp. From my experiments, the greater problem = low and high frequency parasitic oscillations.  Delightfully, I hear and measure 0 hum in his project.

Low impedance Virtual Ground [VG]

On all 3 boards, rather than run the classic R1= R2 voltage divider, I opted to run a well bypassed, unity gain, op-amp voltage follower to generate the VG. The output impedance = near zero. This may represent overkill, however, as always, you're in charge.

An EE who developed a neuromuscular illness and could no long build gave me a free box of new parts including TL082 op-amps, and they, along with 5532s donated by another reader cost me nothing. I applied 1/2 of  the TL082 dual op-amp on both preamplifier boards for VG and left the other half unused.

In general — in a large single-supply project, a variety of currents sum into VG and unless its impedance were practically 0, a changing voltage might arise which could lead to various unwanted feedback loops; both positive and negative. Fortunately, 0 impedance is fairly easy to obtain with an op-amp buffer due to its high open-loop gain at lower frequencies plus low capacitive reactance in the applied bypass capacitors at higher frequencies.


Above  — The typical scheme to deal with unused op-amp halves. For some op-amps, floating inputs doesn't matter; however, in others such as CMOS types, problems may arise. So most people just connect the unused op-amp pins in a fashion like I've shown to remove any worries.

Above  — An Ugly breadboard under construction where only 1 half gets used.

Preamplifier Board One 

 Above  — Preamplifer board 1 schematic.

The 39 Ω series resistor + 220 pF shunt capacitor form an RC network low-pass filter to bypass RF off of the signal path input where it might getting rectified and wreck things. My also RF bypassed signal source ( receiver ) line output impedance will = 39 to 50 Ω, so this RC filter's 3 dB cutoff frequency lies somewhere between 8 - 9 MHz which isn't very good, but so far it's working OK. I'll increase the capacitor value if needed. The die cast chassis helps the cause too.

All op-amps V+ pins are 10 Ω resistor decoupled plus bypassed for both AF and HF per the 5532 datasheet. After the unity gain voltage follower buffer, a 13.1 dB power gain stage ( rolled out as a 4 pole, low Q, low-pass filter) gently scrubs off frequencies above 6 KHz. My listening tests over the past 2 years suggest this fCo works great for general AM, FM and CW listening. I think of it as subtle filtration of the receiver audio without rendering ringing, nor removing the sizzle.

All the signal path resistor values in the preamplifier boards are low to reduce Johnson noise. The low value 270 Ω resistor AC coupled to ground in the low-pass stages might raise your eyebrows, however, measured versus calculated losses caused by the R is only 1-2 dB and I accounted for it when choosing the power gain.

Preamplifer 1 Output

The output goes to a buffer with a "line-out" AC-coupled RCA jack along with a thud removing 22K shunt resistor — plus to a 500 Ω potentiometer for the main audio channel. I wanted the "line output" to view the signal coming out of the first preamplifer to help me determine if distortion occurs @ the input of preamplifier 2 with louder received signals. * Since I know preamplifier 1's gain, I can calculate the signal source output voltage without open a chassis lid. 

I'm not sure what AF signal source amplitude is ideal, and of course with the various receiver modes and RF signal levels seen this might vary widely. Thus, I placed an input gain control after preamplifer 1 to accommodate a wide variety of input signal strengths. The main point = if the signal is too strong I can reduce it to prevent clipping in subsequent stages. Over time, I'll figure out what works best in this application.

* Note: I don't employ AGC nor headphones in my home station receivers.
Preamplifier Board Two

Above  — Preamplifer board 2 schematic.

A 2 pole 6 KHz cut-off low-pass filter boosts the signal 12.6 dB and drives the single knob tone circuit. Adapted from Douglas Self [ Reference 1 ] this amazing circuit delivers quite an impact: relatively low noise compared to some tone stacks, plus low distortion. I'll cover the 20 pF feedback capacitor in the tone control op-amp in Part 2. 

I love a tone circuit in my receiver audio chain. Most receivers lack a tone control circuit. Not knowing the listening environment, nor speaker we'll listen to, we must chose fixed coupling capacitors in our receiver AF chains — and hope for the best. 

Sometimes, we'll choose small value coupling caps like 0.1 µF and get tin cup audio. Conversely, muffled speaker audio may come with too large a coupling capacitor value(s). A tone circuit allows you to compensate to maintain fidelity and also personalizes the listener experience a little.

A simple, low noise FET switch grounds the tone op-amp output for use with a transceiver.

Not shown in a schematic but seen in photos — the DC mains on - off switch is a 1K pot with a integral switch. I may implement a side tone circuit + volume control at some point and placed this part to allow easier future modifications.

Above  —An FFT of the tone circuit when pushed into subtle clipping @ 8.81 Vpp. At 8.64 Vpp, no harmonics were seen ( just the DSO noise floor ). I've never viewed a single 12 VDC supply tone control circuit deliver such headroom before on my bench. Great design by D. Self !

Above  — The preamplifer number 2 breadboard. For space reasons, I placed the FET switch on the AF power amplifier board ( board number 3 ). The 2 squared off islands got drilled when this board was bolted in. All my 2.2 - 4.7 µF polyester caps are rated @ 400-600 V ( a bargain bin purchase from long ago ) so they take up lots of space.

PA Driver and Finals

Above  — Driver and PA schematic. I first presented this PA design in AF Power Amp Experiments and also placed it in Regen #5
I've further refined this design to remove some noise and boost stability.

Setting the PA Bias

To set the bias of the Class AB pair in my PA designs, I first connect a 1 KHz oscillator to the input and terminate the output in an 8 Ω resistive load.  A DMM set in ammeter mode inserted between the power supply and positive rail monitors DC current to help prevent excessive current draw situations such as a breadboard mistake. The signal generator input signal is initially set to 0 via it's amplitude control pot.

I initially set the entire stage quiescent current to 25-30 mA by adjusting the VBE multiplier "amplified diode" 10K trimmer pot. Then the signal generator pot is adjusted to until about 5 Vpp appears in the DSO. The VBE trimmer pot is tweaked to just eliminate crossover distortion in the time domain sine wave. While slowing increasing the signal generator amplitude I then find the maximum clean average power and ensure that crossover distortion doesn't re-emerge with more drive. Maximum clean Vpp measures show between 9 and 10 Vpp for this PA. 

After setting the bias and determining maximum clean power, reduce the signal generator amplitude to 0 and view the current meter. This is the entire stage quiescent current and typically runs 20-50 mA for my PA. I then remove the ammeter, repeat the whole process without the DMM 2 or 3 times. Finally when satisfied all is OK, the quiescent current gets re-measured and recorded. 

I set the bias in my homebrew audio PAs by viewing in time domain for decades — until a fine DSO landed on my bench.

As mentioned in previous posts — if you've got a DSO with a good FFT math function, another  way to set the bias = tweaking the VBE multiplier pot while watching the distortion in your FFT sweep. That's what I do.  My DSO features 12 bit sampling in high resolution mode. In this particular amp, I set the bias pot for the lowest distortion performance, turned off the signal generator and then measured the entire stage quiescent current @ 45 mA.

Above  — PA bias time domain screen shot. I advanced the VBE pot just a fraction to eliminate this crossover distortion. Then I added in FFT to really nail the bias setting by repeatedly taking it just in — and then just out of crossover distortion to find the sweet spot with the lowest measured distortion.

Above  — PA driven by signal generator to 1.49W: some clipping emerges and notice it's happily ~symmetrical on both halves as the signal swings nicely 'tween the rails.

Above  — PA breadboard on circuit board number 3. I'm increasingly using AF + RF capacitors as stand-offs in my Classic Ugly Construction.  I didn't need to separate the PNP from the NPN transistor as shown above in the Cu board heat sink, since, in my design, they 're connected anyways. A carved cut separates the heat sink collector connection nodes from the copper ground plane board.

Above  — PA breadboard alternate view. Right after, I added the op-amp circuitry and fired it up.


[1] Power

Above  — FFT showing my "clean" average power = 1.38 W. My personal definition of clean average power is where all harmonics are very close to 60 dB down. Use your own definition please.

Above  — FFT showing the harmonics where Vpp = 6.45.  In a quiet room, 650 mW is loud to my ears. I don't use AGC when receiving at home and prefer no clipping on strong signal peaks so low distortion = desirable in my world.  Headroom happiness.

[2] Output Noise Voltage

Above  — An averaged measure of noise across a 8 Ω resistive load with no input signal and the input port terminated with a shunt 470 Ω resistor. No oscillations and low noise voltage make me happy.

In general:  

For a good amplifier,  the noise level (over the audio bandwidth) should be at least 80 dB below the maximum output power — preferably it should be >= 90 dB. For 1.38 watts and 8 ohms, that's 3.3 Vrms. The noise (bandwidth limited to audio) should then be less than 330 µVrms; or about 2 mVpp maximum.  You can see I made it.

I'll go into debugging in Part 2, but the key for a very low output noise voltage in my particular amp = the 56 pF feedback cap from the op-amp output to input + the 47 pF cap across the 27 K feedback resistor that limit the PA stage's upper bandwidth.

In the highest spirit of the grunt bench experimenter, I tried various cap values and recorded the noise voltage results. The capacitor value across the 27K feedback R was crucial. I had to keep it low. In this amp topology, even a 270 pF capacitor in that slot could trigger intense HF oscillations. More later.


I prefer measures, but noticed a correlation: good measures often leads to good sound in AF design work. Easily the best AF I've built, this box lacks noise plus distortion and sounds wonderful  — Hi Fi bench amplifier goal accomplished.

Miscellaneous Photos

Please click on a photo to page through them...

Above  —  The old lab AF amp dwarfed by the new version. You can find it in the old site archive. I used this old blue box for speaker AF listening tests of the preamplifier 1 and 2 breadboards.


Above  —  The AF amp lying on top of my 4 linear bench power supplies. 3 are home brew. I modified the commercial product (bottom right) to improve its performance —  I use it in QRP-level transmitter design because of a big transformer that makes it a stout voltage source.

I'm not the only fan of my big new bench.


[1] Small Signal Audio Design: 2nd edition by Douglas Self. I briefly review this amazing book and stellar author on my Funster Part 2 blog post.

I'm not 1 for hyperbole, but a low noise signal generator, a 'scope and Douglas Self's book are all you really need to design good AF circuitry on your bench.

[2] Professor Kenneth Kuhn:  This page and also as a mentor of bench excellence.

[3] EMRFD published by the ARRL:   In EMRFD, Rick, KK7B inspired me to passionately pursue AF design. Thanks to Wes and he.

[4] Elliott Sound Products -  Rod Elliot's pages = essential reading for AF designers. 

[5]  Ask The Applications Engineer - # 25 from Analog Dialogue Q & A [ Analog Devices ]

Click for Part 2