Tuesday, 11 November 2014

Measuring Resonator Q at VHF

Greetings!

I've long enjoyed broadcast FM DX chasing. This post covers the first chapter in my pursuit to design and make a homebrew broadcast (wideband) FM DX receiver.

Traditional FM DX superheterodyne receiver ran single conversion with a 10.7 MHz IF and usually a dual-gate MOSFET mixer behind 1-2 dual gate MOSFET preamplifers embedded in 2-4 doubled-tuned bandpass filters.  To cover 88-108 MHz with decent skirts + bandwidth, the band-pass filters were tuned with ganged, air-variable capacitors that simultaneously tuned the VFO.

Fast forward to this day in time. Following low-pass filters and T/R switch circuitry, some modern narrow band FM transceivers run varactor tuned band-pass filters, 1 or 2 dual gate MOSFET amplifier(s) and a MOSFET mixer in the receiver chain.

I explored varactor tuned band-pass filters for wideband FM and learned a few things along the way.


Above — My first FM band-pass filter that tuned all 20 MHz of the FM band without a double humped filter response. In order to get the BB639's capacitance down to its lowest value (~ 3 pF to tune 108 MHz), you need 28 volts applied reverse DC. I keep a homebrew bench DC-DC converter for just that purpose. The filter's 3 dB bandwidth ran from about 2.2 MHz at 88 MHz to 3.3 MHz wide at 108 MHz

In order to boost the filter low-pass response, I tapped the inductors for the input and output ports. Light loading improved the filter response, but reduced the input/output return loss. Tuning a filter over a 20 MHz span proved a lesson in compromise.

60 nH isn't a lot of L and my coils were 6 turns of bare copper 22 gauge wire wound on a small bolt and then stretched to allow room for tapping and to set the correct measured inductance. I further tweaked them in-situ. Later, I placed this filter after various preamplifiers including a dual-gate MOSFET and a common gate JFET amp.


Above — A snippet of 1 of my amplifier + band-pass filter circuits. I got a better filter response with a 96 nH inductor, although this should technically worsen it.

1 major problem arose with my filters: a horrible insertion loss of 8-9 dB!!  I expected about half that. Later I wrote some great friends for advice and after reading their wisdom, I came up with 2 sane theories: the insertion loss was due to lowered Q and input + output port mismatch.

Resonator Q Measurement

I then realized that I'd never measured resonator Q at VHF. If you own a Q meter such as the
HP4342a stop reading now, get a coffee and go to another blog-site.

As amateur experimenters, to derive resonator Q we may employ 2 techniques: calculate Q after measures with a parallel tuned L C tank, or calculate Q after measuring with the L and C set in a series tuned trap circuit.

I've not enjoyed much success with the latter, so will present the method where our parallel L and C are loosely coupled to a 50 Ω source and load. Please refer to EMRFD page 7.36 for more details.

Supplies:
  • Signal generator with level output amplitude,  50 Ω output Z and enough output power to allow measurement with your particular detector.
  • 50 Ω detector: spectrum analyzer,  50 Ω terminated 'scope, or measurement receiver etc.
  • 50 Ω patch cables.
  • A homebrew jig with RF connectors, coupling capacitors and ground plane.
  • Frequency counter.
  • Through connector.

Setup:


Above — The basic paralleled tuned resonator measurement set up including gimmick probes as the input and output coupling capacitors.



Above — My test jig with the inductor and air variable capacitor soldered in place. I copied this jig from Bob, K3NHI and received advice from Wes, W7ZOI. Bob made a circuit at his QTH, measured the resonator Q and sent me a photo and some measures by email. I re-created his circuit to compare results. The coil = ~ 300 nH or 10 turns on a 1/4 inch bolt. The capacitor to resonate it @ ~100 MHz on my bench = 7.27 pF.

 Method to Set Jig Insertion Loss at VHF

In order to measure a resonators unloaded Q, or Qu, the insertion loss of the jig must minimally be 30 dB. To clarify, join the input and output cables together with a barrel or through connector and measure power. Unsplice and then connect the cables to the jig and measure power once again — power should drop by at least 30 dB at the test frequency.

To create the >= 30 dB insertion loss we lightly couple the jigs input and output with low value capacitors. At HF, we may insert small series capacitors, but this is nearly impossible at VHF unless you own some special microwave parts. Instead of series capacitors, we couple with gimmick wires.  Experiment to find the correct wire distance from the resonator to create the needed insertion loss.

Here are my 2 jig measurements with a DSO plus the IL calculation:





Above — My insertion loss fulfills the require >= 30 dB needed for proper resonator coupling
I calculated the IL in dB with JavaScript Tool G. It's difficult to measure under 50 mV with a 'scope for some; your power meter or spectrum analyzer might work better.

To get the needed IL or measure for Qu calculations, set your signal generator to the desired frequency and then tweak the resonator variable capacitor to give the highest possible AC voltage/power. Then re-tweak your signal generator to ensure you've peaked the signal. You might have to re- tweak the variable cap again and so forth.

Optionally, If your resonator capacitor is fixed, adjust your signal generator frequency to peak the signal.

After properly setting the IL and peaking for the strongest signal, the final measures go quickly:

  1. Record the frequency where you measured maximum power: that's FO, or center frequency.

  2. While watching your detector, lower the signal generator frequency until FO power drops by 3 dB [easier to do with an SA or power meter]. Record that frequency.

  3. Bring the signal generator back up to FO and then increase frequency until the power drops by 3 dB. Record that value.
  4. Calculate Qu as Frequency/Bandwidth.
I'll show my measures performed with a homebrew signal generator with less than ideal tuning resolution; however, you'll get the idea.



Above — FO or center frequency = -25.84 dBm, therefore my 3 dB down target = 28.84 dBm when I change my signal generator below and above FO.



Above 2 images — The 3 dB measures below and above FO (getting as close as possible with a homebrew VCO).  My VHF VCO sports a >= 30 dB output return loss from 98 - 149 MHz.
Calculate B or the 3 dB bandwidth by subtracting the lower frequency from the higher.


My calculated resonator Qu = 289. Bob, K3NHI measured then calculated 300 on his. Pretty close.

Bob's homebrew jig and resonator.

Above — Bob's homebrew jig and resonator. Bob measures everything: including his breadboard length! 

Varactor Measurement

I removed the air-variable trimmer cap and inserted a small piece of copper clad break-out board to hold a tiny BB639 varactor (size SOD-323). I voltage tuned it to resonance and then repeated the whole resonator Q measurement routine. Q = 174: a drop of 115 which would boost my original FM band-pass filter insertion loss by at least 1.5 dB compared to an air variable capacitor.


Measuring resonator Q with a varactor.

Above — Measuring resonator Q with a varactor.

Through experiments with the the aforementioned parallel + series resonator measurement techniques, I learned that carved squares and traces in boards may also lower Q. Even a Manhattan or carved pad nearby may couple to the resonator and drop its Q during measurement. My worse case measure produced a drop in Q of 20 from nearby islands carved in the copper board. Clearly we need board traces, but they can affect resonator Q and thus add to filter insertion loss.

Further, good VHF filter designs stick each resonator in an RF tight compartment. Whatever filter I eventually keep, I'll mind my Qs.


Footnotes

Thanks to Bob, Bob, Wes, Ken, John and others who kept me on track  — I know just enough to act foolishly on the bench.

I made 1 varactor tuned band-pass filter on single-sided copper board and compared it to the double-sided board versions. The single-sided board suffered poor stopband shape and didn't tune as well.

double clad board


Above — A board set up for a dual-gate MOSFET surface mount circuit where the FET source runs a shunt resistor and capacitor to ground, plus has DC voltage on both G1 and G2. Some via holes connect the top and bottom ground plane.

double clad board


Above — I placed 22 gauge copper wire in the via holes and soldered them top and bottom.
Grounded parts are placed near a via wire. Sometimes, I'll add more via wires near grounded parts.

100 MHz amp

Above — A single frequency amplifier carefully matched to see how much gain the BF998 could deliver. 21.2 dB rocks my world.

Some of my (mostly) 50 Ω homebrew bench modules for test and measurement

Above — Some of my (mostly) 50 Ω homebrew bench modules for test and measurement.



return loss bridge

Above — My favorite design project of 2014: a return loss bridge with directivity >= 30 dB from 5 MHz to 1.5 GHz.   You may read more about it in the old site pops.net archive: Topics 2012 - 2014 : Caitlyn 310 — UHF Beginnings : 3. Return Loss Bridge Experiments : Bridge #4



Saturday, 8 November 2014

AF Power Amp Experiments


Although VHF focused, I spent some time studying AF power amp design this fall. Even SDRs need audio amplifiers for the ear interface.  If a speaker guy wants some serious wattage blaring in his cottage, then a split DC supply audio power amp with DC -15v / +15v or so garners an easy ~12-16 watts clean average power.

However, back in the land of radio heads, swinging the AC even remotely near rail to rail with a 12 volt single-supply proves arduous. Thus, designing and building AF power amps that cleanly swing as close to the supply rails as possible seems like a good idea.
Before these, my single-supply AF power amplifier experiments usually employed single or Darlington emitter followers arranged as a complimentary symmetry pair. 1 bleak fall day; with coffee and cats, I changed it up and moved away from typical, symmetrical, emitter follower push-pull finals to learn about other topologies.

I tested these amps with an ultra-low distortion, variable amplitude 1 KHz signal generator, an 8 Ω resistive dummy load, a DSO with FFT and 2 DMMs. I'll share 3 progressive amp experiments:

Amplifier 1

First up is quasi-complementary amplifier. Unusual to see in 2014, but great fodder for learning what goes on in a PA.  After making, debugging and analyzing these circuits,  I felt humbled about how little I consider AC signal phase in addition to the easier concepts of amplitude and frequency during my AC signal analysis.

My first quasi-complementary amplifier.

Above in Figure 1 — My first quasi-complementary amplifier.

Q1 = a class A driver with negative feedback that lowers its input impedance. Typically, a small series resistor goes on the input to raise its input impedance, however, I left it off on this experimental amp. 

Q2, an emitter follower [with no phase inversion] forms a standard Darlington pair with Q4 Their asymmetrical compliment are Q3 and Q5: Q3 is a PNP common emitter amp that inverts the signal phase.

When a negative swinging signal enters Q4, it draws current that flows into the speaker because simultaneously, a positive swinging signal enters Q5 and holds it near cut off. When the AC signal flips polarity, Q4 cuts off and Q5 conducts. Thus the quasi-complementary amplifier gives single-ended push-pull output.

Let's discuss applying forward bias on Q4 and Q5 to eliminate crossover distortion.

The simplest method involves placing diode(s) across the Q2 and Q3 bases to forward bias them enough to in turn, forward bias Q4 and Q5 almost to conduction in their quiescent state.


Above — Cross over distortion measured across the 8 Ω load with 2 bias diodes.



Above — Cross over distortion with 3 diodes (quite near to eliminating the cross over distortion).

I could have tried 4 diodes, but just replaced the diodes with an adjustable level shifter or so-called amplified diode to allow precise control of the forward bias on Q4 and Q5. Further, 1 replaced the original DC coupled feedback from the speaker back to the Q1 with AC coupled negative feedback (a 100 pF capacitor):

Figure 1B. My final quasi-complementary amplifier.

Above — Figure 1B. My final quasi-complementary amplifier. 

The 22 µF capacitor in both designs provides essential positive feedback across the 1K resistor. Bootstrapping feedback compensates for the asymmetrical output stage allowing the positive peak signal swing to approach its negative counterpart. The 22 µF capacitor maintains a charge to keep the DC voltage across R1 constant and their time constants must consider the lowest frequency to be amplified.

Figure 1B breadboard;


Above — Figure 1B breadboard; although it still has diodes to set the quiescent current at this point.


Above — FFT  in pink showing strong harmonics. You can see heavy cross-over distortion in the yellow time domain tracing too. I then tweaked the 10K level shifter bias pot to eliminate the crossover distortion:


FFT (pink) with minimal cross-over distortion after adjusting the level shifter pot.

 Above — FFT (pink) with minimal cross-over distortion after adjusting the level shifter pot. 

I'm now routinely setting my level shifter to remove cross over distortion with the aid of FFT plus simultaneous viewing of the signal in time domain ('scope viewing). I usually set the drive so the amp is making about 1/2 its maximal clean signal power when tweaking the bias. Then I increase the drive to "full" clean signal power to confirm that cross over distortion doesn't re-emerge, Usually at this point, harmonic distortion begins to dominate.

While it's possible to set the PA forward bias by just viewing the sine wave and tweaking the 10K pot to find where the cross over distortion just disappears, to me, the FFT takes this to the next level. 

My Figure 1B bench assessment offered a good new-bad news paradox. In the 'scope trace above, the tones are down 63 dBc, so I managed my personal best, single DC supply PA in terms of distortion at 1/2 power. The bad news = it took ~ 200 mA of quiescent current to deliver this performance. 

It seems that the output is stage is starved for gain on the upper half of the AC cycle — which is why it takes so much current and it's peak-peak is limited. Point X, or the collector of Q5 should ideally be close to 1/2 VCC, but in my quest for low distortion and big signal swing, it ended up at 5.68 VDC.

I think back in the day when they used amps like this, builders tolerated a lot more crossover distortion, but I’m surprised that this amp when biased in AB (or maybe deeper) gives such good distortion performance with the mere addition of that 100 pF feedback cap. Further, I liked setting the PA bias with the aid of FFT instead of just looking at the old sine wave.  I'm learning, and for sure, making mistakes.


Above — FFT of Figure 1B's maximal clean signal power = 3.82 Vpp [228 mW]. The worst tone is -53 dBc. Clearly with ~ 200 mA of idle current and only 228 mW clean signal power, this power amplifier is not a keeper, but I enjoyed analyzing + working on it and felt a boost in confidence going forward.

Signal Squaring

A home brew square wave board might help assess your AF power stages in time domain. If your PA can accurately reproduce a square wave, you're on the right track!

A square wave provides a symmetrical waveform that alternates instantaneously between 2 levels and allows you to see rise times, overshoot and other phenomena. I used square wave analysis to help me choose the 100 pF AC feedback capacitor in Figure 1b.

Ensure you watch your DC current so you don't suffer final amplifier thermal runaway during square wave testing as your amp may consume significant power when driven hard. I blew 1 final in my experiments. With PA experiments, heat sinking the finals proves worthwhile. I did this in Amplifiers 2 and 3.


Above — A simple, signal squarer I keep in my lab. With appropriate RF bypass and series coupling caps  it also works great at radio frequency. For example, 0.1 µF capacitors @ HF.


A ragtag squarer breadboard that's seen much use over the past few years.


Above — A ragtag squarer breadboard that's seen much use over the past few years.

Above — Square wave output of Figure 1b

Above — Square wave output of Figure 1b.

Amplifier 2

I converted Amplifier 1B into a full complimentary version:


Above — Figure 2 schematic. A Darlington — complimentary circuit lies on both halves of the push-pull scheme, Function is similar to Figure 1B, but the symmetry adds the advantage of a remarkably low quiescent current for proper AC/DC operation. I added emitter degeneration (series feedback) to the Q1 voltage amplifier. Nearly every resistor was adjusted or swapped to find the sweet spot in order to swing the largest AC voltage 'tween the rails.

Breadboard of Figure 2 connected to an 8 Ω resistive load (2 parallel resistors).

Above — Breadboard of Figure 2 connected to an 8 Ω resistive load (2 parallel resistors).

Bias setup

Above — Bias setup: Even at half-power, the lowest distortion possible gives a 2nd harmonic of only 32 dB down. I tried many schemes to lower this distortion, but failed.


Above —Maximum clean signal power = a disappointment. Even with lower drive, this amp suffered from harmonic distortion. No point in continuing to work on it. Onto Amplifier 3:

Amplifier 3

In review; Figure 2 featured a complementary NPN—PNP driver + PNP—NPN output pair with a level-shifter that sets the bias differential on the drivers which in turn establish the bias for the finals since they're directly coupled. Sadly, Figure 2 suffers from distortion and a high quiescent current.

Amplifier 3 fixes these woes:

Figure 3 schematic.

Above — Figure 3 schematic. Optimized for low distortion and quiescent current @ 1W; it's driven with a low noise op-amp and ranks as the best single-supply audio PA I've built to date. 

A combined time and frequency plot at maximum clean power: 1 Watt. The 2nd harmonic is 63 dB down!

Above — A combined time and frequency plot at maximum clean power: 1 Watt. The 2nd harmonic is 63 dB down!  

Following FFT analysis, I performed the most important test of all — listening through a speaker. 

On my bench top sits an old cassette player with line-level output. The audio tape plays the clear — booming — voice of a loud, male Russian professor. In Russian language, only 1 syllable is ever accented and his taped voice peaks rip like thunder — well testing audio amps plus scaring cats. Wow, классный , superb --- for sure, a version is going in my next receiver.

When you connect an op-amp to an AF power stage expect oscillations. Surprisingly, mine were between ~1 and 2.7 MHz. With the output connected directly to the op-amps inverting input you might create a situation where the total phase shift at the feedback loop exceeds 360 degrees plus exhibits a gain > 1  — oscillations — 

Experimenting with all the parts connected to the op-amp's inverting input is the first place to start when debugging higher frequency oscillations. 

Decoupling and bypassing the DC supply with simple RC networks is warranted if motor boating [low frequency oscillations ] arise in any AF circuit.  On my JavaScript RF Tools page,  Section D. Calculate Cut off Frequency for an RC Pi Low-Pass Filter , you may assess combinations of caps and resistors.

Figure 3 breadboard shown in audio test mode

Above — Figure 3 breadboard shown in audio test mode with an RCA female on the output and a "tacked on" 10K volume pot since the voltage gain is pretty high. This Cu board is the same 1 used for Figure 2, so by this time it's looking pretty worn. 

I think some radio enthusiasts feel more impressed by pretty looking circuits rather than well designed and properly tested stages... I thought I got a good exposure though. I have never used Photoshop and prefer to get it done on camera.

Also, we normally don't leave unused op-amp pins floating — I will have stripped parts and trashed this board by the time you read this.

Final Thoughts

I've got some other ideas, designs and advice to assess. Further, some rail-to-rail op-amps will arrive soon. In 2015, I'll see if I can make a linear PA without distortion at signal amplitudes even closer to the rails. Perhaps? Many thanks to my mentors + supporters and

Thanks to you for reading — catch you later!


Wednesday, 29 October 2014

Harmonic VXO


Greetings:

I needed a quick, very temperature + frequency stable oscillator @ 50.8 MHz to test a receive mixer. Already running elsewhere on my bench, my main VHF signal generator wasn't an option.

Without going to a digital signal box, I weighed my build options:

  • VFO — difficult to stabilize at VHF.
  • VCO — even more difficult to stabilize at VHF and it's too much work to make a PLL for it.
  • Overtone crystal oscillator — little chance of finding a crystal to vibrate at 50.8 MHz.
  • Fixed crystal oscillator  — need a crystal to vibrate at exactly 50.8 MHz. Right!
  • VXO  — do I own a 10.16, 16.93 , 25.4 MHz etc. crystal ?
  • VXCO — still need a crystal to double, or take a harmonic from. The varactor will add phase noise, but might work.

Then came a mad search. Soon enough, I found a long forgotten bag of crystals and after 10 minutes of fumbling with it, I felt jubilant to find a 16.9344 MHz crystal!  Good, I'll make a VXO with an output filter to help suppress all but the 3rd harmonic.

I grabbed a copper board and began soldering — no need for a schematic, for I've made many tens of xtal oscillators in my sleep. About 20 minutes later, I was done and ready to measure the ugly thing.

Breadboard

At ~50 MHz, I foolishly went with single-sided Cu board and apart from a couple of carved squares, built with super-fast, classique Ugly Construction. The yellow toroid core = a T50-6, while the black parts are T37-10 powdered-iron cores. Low cost, low Q, garbage-grade trimmer caps employed.

A harmonic fixed xtal oscillator come VXO

Above — A harmonic fixed xtal oscillator come VXO. First I ensured that the 16.9344 MHz was indeed a fundamental crystal by shorting the transistor's collector coil and measuring with a 10X probe ---- DSO with a built in frequency counter.

The doubled-tuned circuit values were chosen based upon my work with 6 Meter band-pass filters 2 years ago. As I recall, the 3 dB bandwidth is around 1.8 MHz for ~250 nH inductors, a tuning capacitance around 31 pF and a 1 pF coupling cap between the resonators.

I build progressively: First made a fixed crystal oscillator and then after tuning up the output filter, measuring DC + AC voltages and current, I then added the VXO parts shown below the red arrow. The 3.8 µH inductor was just something lying in my #6 material toroid parts drawer.

I keep several pre-wound inductors in each size hand to speed up design work. All I have to do is remove a few turns as needed, or go to a toroid that looks like it contains more wire. I don't count turns except when making transformers and instead rely on an LC meter to measure the inductance.

Spectrum analysis after tuning the VXO for ~50.8 MHz

Above — Spectrum analysis after tuning the VXO for ~50.8 MHz.  I red numbered the tones starting at the crystal's fundamental frequency. The strongest harmonic = the 4th tone @67.6 MHz lying  -43.9 dBc.

The VXO  output in time domain

Above — the VXO output in time domain.

A triple tuned version.

Above — A triple tuned version.

The double tuned version contained too much harmonic energy, so I added another identical tank and re-measured.

Spectrum analysis of the TT harmonic VXO.

Above — Spectrum analysis of the TT harmonic VXO. Strongest tone = -56.7 dBc; most are under 63 dB down.

Now that's what I'm talking about!  Onto my mixer experiments...

73,  Todd---VE7BPO---

Friday, 24 October 2014

Funster Receiver Notes Part 3

Funster Receiver Notes Part 3
Audio PA, side tone and wrap-up.

The most common home brew receiver part = the LM386 audio amplifier.  A Signetics brainchild like the NE/SA602, the LM386 first appeared in home brew receivers in the late 1970s. The impact of the Signetics design team on modern home brew radio building exemplifies an outward phenomenon.

Although, I too occasionally drive speakers with the old 386, distortion develops at just ~300 mW average power and nobody's ever called it a low-noise part; especially when the internal gain is set higher than 20.

For years, I've plodded to find a popcorn discrete replacement for the LM386. In the Funster receiver, I evolved my basic popcorn AF PA design a little more — it still needs work, but I can look you in the eye and tell you that Funster sounds great and exhibits low AF chain noise.

My earlier popcorn AF power amps ran low voltage gain and I didn't realize this was a problem until readers emailed to say so. I get it now — in our RF home brew community some expect the full-on gain (200) of the LM386!  In contrast, when I run a 386, I set a gain of 20-50. Talk about different perspectives. That's what makes home brew RF design exciting.

Popcorn Audio Power Amplifier


My first popcorn AF stage with a 5532 non-inverting amplifier

Above — My base popcorn AF stage with a 5532 non-inverting amplifier features adjustable gain to appease builders who rely on the PA stage for most of their voltage gain. You may change the fixed and/or trimmer series resistors between pins 1 and 2 to set your desired gain either by listening to your receiver, or crunching some op-amp arithmetic [ Vo = Vin (1 + R2/R1) ]. I fixed the other half of the 5532 op-amp as a voltage follower / rail splitter.

The 2 power followers pairs are biased into Class AB by a 2N4401 level shifter. Tweaking the 10K trimmer pot even the slightest may change your idling or quiescent current dramatically; so carefully set the bias with your test equipment switched on.

I wrote a web page on biasing your AF finals: 2006 - 2009: Complimentary Symmetry Amplifier Biasing Basics.  You'll find it in the Old Site archive. A quick review follows:


Measuring quiescent current + the voltage dropped across both NPN/PNP power follower bases.
 
Above — Measuring quiescent current + the DC voltage dropped across both NPN/PNP power follower bases. Typical popcorn AF amplifier values are shown. In most cases, you'll measure 1.1 to 1.4 VDC across the final pair with properly set bias. 

I set the final amplifier pair bias in the popcorn AF stage just like I used to with 100 watt guitar amplifiers: put an appropriate resistive load on the output, connect a signal generator to the input and tweak a pot to find the set point where the crossover distortion disappears.

I keep 2 DMMs on my bench: 1 serves exclusively for power measurement and I never fire up a "just built" power amplifier stage of any sort without an ammeter connected between the DC power supply and VCC point.  I only want to enjoy my measurement experiences — the ammeter will catch any shorts or other problems long before thermal runaway takes out your NPN power transistor (we rarely blow the PNP in a complimentary pair).

With no applied signal, your ammeter will read the quiescent current of your whole PA stage (some purists argue that quiescent current only applies to the final complimentary pair). I usually set the bias and measure my quiescent current after the finals have warmed up. It's best to thermally couple the adjustable NPN level shifter to the PA heat sink, but you may omit this in popcorn-class PA stages.

After setting the bias, switch off the signal source. After that, measure across the 2 BJT pair bases and then look to see what the ammeter shows. Expect 20-35 mA quiescent for a warmed up popcorn amplifier.


Above — Average power measurement and formula. A 1 KHz low distortion signal generator makes a great weekend project — mine feature variable gain Wien bridge oscillators with 2-4 poles of low-pass filtration using extremely low-noise op-amps. Without solid measurement tools we're just bench lackeys.

Back to the popcorn power amp shown earlier: 1 strength is that the sections labelled PA (the 1 µF and all the parts to its right) will add a PA stage to any voltage amp with a low output impedance. 1 weakness = the output is open loop — with no negative feedback to reduce distortion. 

In previous experiments, I left out the PA's 1 µF coupling capacitor plus the 4K7 resistors and just drove the power followers with a DC coupled op-amp output. This works fine for headphone-level output power — but I'm a speaker guy. When driving a speaker loudly during signal peaks, the drive to the followers may poop out and distort the signal with a glitch that resembles crossover distortion. Increasing the quiescent current won't fix the problem — I've tried that.


Above — Distortion viewed when swinging a Vpeak-peak of 7v with DC coupled op-amp drive.

Evolution 1.
Above — Evolution 1. I removed the 1 µF coupling capacitor (good riddance) and arranged the top 4K7 resistor to provide positive feedback. The op-amps were able to drive the followers all the way to normal harmonic clipping with no aforementioned glitch. The bootstrapping also boosted the maximum clean sine wave power.  

Evolution 2. This final PA stage went into my Funster

Above — Evolution 2. This final PA stage went into my Funster. I added a negative feedback loop via a 100 K resistor to put the op-amp into the loop. Do not put a parallel capacitor across the 100K feedback resistor as this will increase distortion in my positive + negative feedback arrangement.

Further, while listening to a variety of CW signals into a speaker. I tweaked the 5532 gain trimmer between Pins 1 and 2 as I adjusted Funster's 500 Ω AF gain pot. When satisfied with the non-inverting amplifier's gain, I removed the trimmer + fixed R between Pins 1 and 2 and measured a series resistance of 21.4K Ω . I soldered in a 22K resistor, retested and felt it gave the prefect amount of PA stage gain for my particular Funster. Let's get to the PA power measures:

The maximum peak-peak voltage of the final Funster PA design into a 8 Ω resistor load = 7.64 Vpp.

Above —The maximum peak-peak voltage of the final Funster PA design into a 8 Ω resistor load = 7.64 Vpeak-peak. To calculate average power we use the peak AC voltage, so divide Vpeak-peak by 2 to get Vpeak. Therefore Vpeak =  3.82v. 

So my clean signal (average) power = 3.82v * 3.82v / 16Ω = 912 mW.

If I increased the drive on my 1KHz signal source any more, the AC signal began to clip. I pushed it just into clipping and then backed off until I eyeballed a pure since wave and got the Vpeak-peak = 7.64 shown above. Admittedly, eyeballing the sine wave feels subjective, however, if you lack a distortion analyzer, a DSO with a good FFT or a sound card/computer audio analysis program, sine wave signal viewing works okay.


Above — The FFT of the sine wave signal above on my DSO showing harmonic tones to the right of the fundamental 1.012 KHz signal. The second harmonic lies 58 dB down indicating my eyeball sine wave assessment works okay. FFT measurement is a better idea though.

Breadboard of the popcorn AF amp I bolted in the Funster receiver

Above — Breadboard of the final popcorn AF amp I bolted in the Funster receiver. I added temporary RCA jacks and tested it on my workbench by listening to Funster.  A shielded cable temporarily connected the installed tone/mute circuit to the rear panel RCA speaker jack. From the speaker jack I patched a shielded RCA cable to the PA input and connected the PA output to an 8 Ω speaker.

This breadboard shown ran a 220 pF cap across the 100K negative feedback resistor — I promptly clipped it out since the capacitor generated distortion.

I'll keep working on my popcorn AF power amp, but this 1 sounds good in Funster and certainly beats the old 386.

Keyed Side Tone


 Above — A phase shift oscillator circuit sent to me by Wes, W7ZOI awhile ago.

I love Wes' side tone circuits and this sine wave generator combined with the mute circuit sounds makes Funster sound like a professional transceiver as I key the companion Funster transmitter.

For amplitude adjustment, I added the 25K trimmer to get a sine wave in your 'scope. I measured and replaced the trimmer with a nearest standard-value resistor in my final build. Many sine wave AF oscillators don't fare well with downstream changes so I added an emitter follower with 4 mA emitter current + AC coupled a 100K resistor on either side of the 10K volume control.

The 100K resistors, attenuate, isolate and add some low-pass filtration to the side tone output. The last 100K R connects to Pin 3 of the 5532 in the PA.

To "key" the circuit, ground the cold end of the 100 Ω resistor with some solid state switch or a key if you want to use this as a stand-alone code practice oscillator. For the latter, don't forget to add some key shaping to remove bounce-bounce clicks. Lacking the correct part, I substituted a 0.27 µF polyester cap for the 0.22 µF called for in the schematic and still pulled off a low distortion sine wave @ ~782 Hz.  



  Above — The side tone signal in my DSO measured with a 10X probe.

AF stage breadboard with the side tone circuit added.

Above — AF stage breadboard with the side tone circuit added.

Wrap-up

The Funster proves a relatively-simple DC receiver offering significant improvement over the standard direct conversion receiver with no opposite sideband suppression. It compares well to the classic superheterodyne receiver that employed a single crystal filter plus front panel phasing control to provide a single, deep notch on the opposite sideband.

As ever, by the time I complete a radio, I would have made it differently. Ours is a hobby where we make our circuits better over time with our test equipment switched on.

Funster Photos (click on the photos to magnify as usual)

Funster Receiver 2014


Funster Receiver 2014 --- front view




Funster Receiver 2014 --- top view



Funster Receiver 2014 --- rear view



A close-up of the NXP BCX56 + BCX53 mounted in a prototype PA board


Above — A close-up of the NXP BCX56 + BCX53 mounted in a prototype PA board. I ran no heat sink other than PC board traces for the collectors. A better choice might be the related BCP56 + BCP53 pair in SOT223 since the bigger package of this BJT pair better sinks heat.
A good through hole substitute = the BD139/140 complimentary pair.

Большое спасибо to all of my mentors and helpers for your support in addition to the 2 workbench companions shown below:

The girlz


I'll show the second simple single signal receiver (Toward Minimal Parts) in the future. I'm now working at VHF for awhile.